Fiber optic gyroscope bias modulation amplitude determination with reset means

ABSTRACT

A bias modulation amplitude monitoring system for a rotation sensor having a pair of output signal component detectors to provide magnitudes of the corresponding frequency components in that signal which are used by a component relationship determination arrangement capable of adjusting the output signal to have a selected value in a range of values for a selected rate of rotation thereof, and to correct variations therein, and having a reset arrangement capable of resetting the system if the output signal indicates an unwanted phase modulation amplitude.

Reference is hereby made to an earlier filed co-pending application byP. Dane et al entitled "Fiber Optic Gyroscope" having Ser. No.07/636,305. This co-pending application has been assigned to the sameassignee as the present application.

BACKGROUND OF THE INVENTION

The present invention concerns fiber optic system phase modulators and,more particularly, to arrangements for accommodating such phasemodulation of electromagnetic waves traveling therein in changingconditions.

Fiber optic gyroscopes are an attractive means with which to senserotation of an object supporting such a gyroscope. Such gyroscopes canbe made quite small and can still be constructed to withstandconsiderable mechanical shock, temperature change, and otherenvironmental extremes. Due to the absence of moving parts, they can benearly maintenance free, and they have the potential of becomingeconomical in cost. They can also be sensitive to low rotation ratesthat can be a problem in other kinds of optical gyroscopes.

A fiber optic gyroscope has a coiled optical fiber wound on a core andabout the axis thereof around which rotation is to be sensed. Theoptical fiber is typical of a length of 100 to 2,000 meters, or so, andis part of a closed optical path in which an electromagnetic wave, orlight wave, is introduced and split into a pair of such waves topropagate in opposite directions through the coil to both ultimatelyimpinge on a photodetector. Rotation about the sensing axis of the core,or the coiled optical fiber, provides an effective optical path lengthincrease in one rotational direction and an optical path length decreasein the other rotational direction for one of these waves. The oppositeresult occurs for rotation in the other direction. Such path lengthdifferences between the waves introduce a phase shift between thesewaves for either rotation direction, i.e. the well-known Sagnac effect.The use of a coiled optical fiber is desirable because the amount ofphase difference shift due to rotation, and so the output signal,depends on the length of the entire optical path through the coiltraversed by the two electromagnetic waves travelling in opposeddirection, and so a large phase difference can be obtained in the longoptical fiber but in the relatively small volume taken by it as a resultof being coiled.

The output current from the photodetector system photodiode, in responseto the oppositely directed traveling electromagnetic waves impingingthereon after passing through the coiled optical fiber, follows a raisedcosine function. That is, the output current depends on the cosine ofthe phase difference between these two waves. Since a cosine function isan even function, such an output function gives no indication as to therelative directions of the phase difference shift, and so no indicationas to the direction of the rotation about the coil axis. In addition,the rate of change of a cosine function near zero phase is very small,and so such an output function provides very low sensitivity for lowrotation rates.

Because of these unsatisfactory characteristics, the phase differencebetween the two oppositely directed traveling electromagnetic waves isusually modulated by placing an optical phase modulator in the opticalpath on one side of the coiled optical fiber. As a result, one of theseoppositely directed propagating waves passes through the modulator onthe way into the coil while the other wave, traversing the coil in theopposite direction, passes through the modulator upon exiting the coil.

In addition, a phase-sensitive detector serving as part of a demodulatorsystem is provided to receive a signal representing the photodetectoroutput current. Both the phase modulator and the phase-sensitivedetector can be operated by a sinusoidal signal generator at theso-called "proper" frequency to reduce or eliminate modulator inducedamplitude modulation, but other waveform types of the same fundamentalfrequency can be used. Other frequencies can be used, and often are, toreduce the frequency level to a more manageable value.

The resulting signal output of the phase-sensitive detector follows asine function, i.e. the output signal depends on the sine of the phasedifference between the two electromagnetic waves impinging on thephotodiode, primarily the phase shift due to rotation about the axis ofthe coil in the absence of occurrence of other significant but unwantedphase shifts. A sine function is an odd function having its maximum rateof change at zero phase shift, and so changes algebraic sign on eitherside of zero phase shift. Hence, the phase-sensitive detector signal canprovide an indication of which direction a rotation is occurring aboutthe axis of the coil, and can provide the maximum rate of change ofsignal value as a function of the rotation rate near a zero rotationrate, i.e. the detector has its maximum sensitivity for phase shiftsnear zero so that its output signal is quite sensitive to low rotationrates. This is possible, of course, only if phase shifts due to othersources, that is, errors, are sufficiently small. In addition, thisoutput signal in these circumstances is very close to being linear atrelatively low rotation rates. Such characteristics for the outputsignal of the phase-sensitive detector are a substantial improvementover the characteristics of the output current of the photodetector.

An example of such a system from the prior art is shown in FIG. 1. Theoptical portion of the system contains several features along theoptical paths to assure that this system is reciprocal, i.e. thatsubstantially identical optical paths occur for each of the oppositelydirected propagating electromagnetic waves except for the specificintroductions of non-reciprocal phase difference shifts, as will bedescribed below. The coiled optical fiber forms a coil, 10, about a coreor spool using a single mode optical fiber wrapped about the axis aroundwhich rotation is to be sensed. The use of a single mode fiber allowsthe paths of the electromagnetic or light waves to be defined uniquely,and further allows the phase fronts of such a guided wave to also bedefined uniquely. This greatly aids maintaining reciprocity as well asthe introduction of non-reciprocal phase shifts as is indicated to bedone below.

In addition, the optical fiber can be so-called polarization-maintainingfiber in that a very significant birefringence is constructed in thefiber so that polarization fluctuations introduced by unavoidablemechanical stresses, by the Faraday effect in magnetic fields, or fromother sources, which could lead to varying phase difference shiftsbetween the counter-propagating waves, become relatively insignificant.Thus, either the high refractive index axis, i.e. the slower propagationaxis, or the low index axis is chosen for propagating theelectromagnetic waves depending on the other optical components in thesystem. In the present system, the slow axis has been chosen in view ofthe optical components used therein.

The electromagnetic waves which propagate in opposite directions throughcoil 10 are provided from an electromagnetic wave source, or lightsource, 11, in FIG. 1. This source is typically a laser diode whichprovides electromagnetic waves, typically in the near-infrared part ofthe spectrum, with a typical wavelength of 830 nm. Source 11 must have ashort coherence length for emitted light to reduce the phase shiftdifference errors between these waves due to Rayleigh and Fresnelscattering at scattering sites in coil 10. Because of the nonlinear Kerreffect in coil 10, different intensities in the two counter propagatingwaves can lead to different phase shifts therebetween. This situationcan be overcome also by use of a short coherence length source forsource 11 which leads to modal phase shift canceling.

Between laser diode 11 and fiber optic coil 10 there is shown an opticalpath arrangement in FIG. 1 formed by the extension of the ends of theoptical fiber forming coil 10 to some optical coupling components whichseparate the overall optical path into several optical path portions. Aportion of the same kind of polarization-maintaining optical fiber as incoil 10 is positioned against laser diode 11 at a point of optimum lightemission therefrom, a point from which it extends to a first opticaldirectional coupler, 12.

Optical directional coupler 12 has light transmission media thereinwhich extend between four ports, two on each end of that media, andwhich are shown on each end of coupler 12 in FIG. 1. One of these portshas the optical fiber extending from laser diode 11 positionedthereagainst. At the other port on the sense end of the opticaldirectional coupler 12 there is shown a further optical fiber positionedthereagainst which extends to be positioned against a photodiode, 13,which is electrically connected to a photodetection system, 14.

Photodiode 13 detects electromagnetic waves, or light waves, impingingthereon from the portion of the optical fiber positioned thereagainstand provides a photo current in response. This photocurrent, asindicated above, in the case of two nearly coherent light wavesimpinging thereon, follows a cosine function in providing a photocurrentoutput which depends on the cosine of the phase difference between sucha pair of substantially coherent light waves. This photovoltaic devicewill operate into a very low impedance to provide the photo currentwhich is a linear function of the impinging radiation, and may typicallybe a p-i-n photodiode.

Optical directional coupler 12 has another optical fiber against a portat the other end thereof which extends to a polarizer, 15. At the otherport on that same side of coupler 12 there is a non-reflectivetermination arrangement, 16, involving another portion of an opticalfiber.

Optical directional coupler 12, in receiving electromagnetic waves, orlight, at any port thereof, transmits such light so that approximatelyhalf thereof appears at each of the two ports of coupler 12 on the endthereof opposite that end having the incoming port. On the other hand,no light is transmitted to the port which is on the same end of coupler12 as is the incoming light port.

Polarizer 15 is used because, even in a single spatial mode fiber, twopolarization modes are possible in light passing through the fiber.Thus, polarizer 15 is provided for the purpose of passing one of thesepolarization modes through the optical fiber, along the slow axisthereof as indicated above, while blocking the other. Polarizer 15,however, does not entirely block light in the one state of polarizationthat it is intended to block. Again, this leads to a smallnon-reciprocity between two oppositely directed travelingelectromagnetic waves passing therethrough and so a small non-reciprocalphase shift difference occurs between them which can vary with theconditions of the environment in which the polarizer is placed. In thisregard, the high birefringence in the optical fiber used again aids inreducing this resulting phase difference, as indicated above.

Polarizer 15 has a port on either end thereof with the lighttransmission medium contained therein positioned therebetween.Positioned against the port on the end thereof opposite that connectedto optical directional coupler 12 is another optical fiber portion whichextends to a further optical bidirectional coupler, 17, which has thesame light transmission properties as does coupler 12.

The port on the same end of coupler 17 from which a port is coupled topolarizer 15 again is connected to a non-reflective terminationarrangement, 18, using a further optical fiber portion. Considering theports on the other end of coupler 17, one is connected to furtheroptical components in the optical path portions extending thereto fromone end of the optical fiber in coil 10. The other port in coupler 17 isdirectly coupled to the remaining end of optical fiber 10. Between coil10 and coupler 17, on the side of coil 10 opposite the directlyconnected side thereof, is provided an optical phase modulator, 19.Optical phase modulator 19 has two ports on either end of thetransmission media contained therein shown on the opposite ends thereofin FIG. 1. The optical fiber from coil 10 is positioned against a portof modulator 19. The optical fiber extending from coupler 17 ispositioned against the other port of modulator 19.

Optical modulator 19 is capable of receiving electrical signals to causeit to introduce a phase difference in light transmitted therethrough bychanging the index of refraction of the transmission medium, ortransmission media, therein to thereby change the optical path length.Such electrical signals are supplied to modulator 19 by a biasmodulation signal generator, 20, providing a sinusoidal voltage outputsignal at a modulation frequency f_(g) that is equal to C₁ sin(ω_(g) t)where ω_(g) is the radian frequency equivalent of the modulationfrequency f_(g). Other suitable periodic waveforms could alternativelybe used.

This completes the description of the optical portion of the system ofFIG. 1 formed along the optical path followed by the electromagneticwaves, or light waves, emitted by source 11. Such electromagnetic wavesare coupled from that source through the optical fiber portion tooptical directional coupler 12. Some of that light entering coupler 12from source 11 is lost in non-reflecting terminating arrangement 16coupled to a port on the opposite end thereof, but the rest of thatlight is transmitted through polarizer 15 to optical directional coupler17.

Coupler 17 serves as a beam-splitting apparatus in which the lightentering the port thereof, received from polarizer 15, splitsapproximately in half with one portion thereof passing out of each ofthe two ports on the opposite ends thereof. Out of one port on theopposite end of coupler 17 an electromagnetic wave passes throughoptical fiber coil 10, modulator 19, and back to coupler 17. There, aportion of this returning light is lost in non-reflective arrangement 18connected to the other port on the polarizer 15 connection end ofcoupler 17, but the rest of that light passes through the other port ofcoupler 17 to polarizer 15 and to coupler 12 where a portion of it istransmitted to photodiode 13. The other part of the light passed frompolarizer 15 to coil 10 leaves the other port on the coil 10 end ofcoupler 17, passes through modulator 19, and optical fiber coil 10 tore-enter coupler 17 and, again, with a portion thereof following thesame path as the other portion to finally impinge on photodiode 13.

As indicated above, photodiode 13 provides an output photocurrent,I_(PD).sbsb.13, proportional to the intensity of the two electromagneticor light waves impinging thereon, and is therefore expected to followthe cosine of the phase difference between these two waves impinging onthat diode as given by the following equation: ##EQU1## This is becausethe current depends on the resulting optical intensity of the twosubstantially coherent waves incident on photodiode 13, an intensitywhich will vary from a peak value of I_(o) to a smaller value dependingon how much constructive or destructive interference occurs between thetwo waves. This interference of waves will change with rotation of thecoiled optical fiber forming coil 10 about its axis as such rotationintroduces a phase difference shift of φ_(R) between the waves. Further,there is an additional variable phase shift introduced in thisphotodiode output current by modulator 19 with an amplitude value ofφ_(m) and which varies as cos(ω_(g) t).

Optical phase modulator 19 is of the kind described above and is used inconjunction with a phase-sensitive detector as part of a demodulationsystem for converting the output signal of photodetection system 14,following a cosine function as indicated above, to a signal following asine function. Following such a sine function provides in that outputsignal, as indicated above, information both as to the rate of rotationand the direction of that rotation about the axis of coil 10.

Thus, the output signal from photodetection system 14, includingphotodiode 13, is provided through an amplifier, 21, where it isamplified and passed through a filter, 22, to such a phase sensitivedetector means, 23. Phase-sensitive detector 23, serving as part of aphase demodulation system, is a well known device. Such aphase-sensitive detector senses a change in the first harmonic, orfundamental frequency, of modulation signal generator 20 to provide anindication of the relative phase of the electromagnetic waves impingingon photodiode 13. This information is provided by phase-sensitivedetector 23 in an output signal following a sine function, that is, thisoutput signal follows the sine of the phase difference between the twoelectromagnetic waves impinging on photodiode 13.

Bias modulator signal generator 20, in modulating the light in theoptical path at the frequency f_(g) described above, also generatesharmonic components in photodetection system 14. Filter 22 is a bandpassfilter which is to pass the modulation frequency component of the outputsignal of photodetector 14, i.e. the first harmonic, after itsamplification by amplifier 21.

In operation, the phase difference changes in the two oppositelydirected propagating electromagnetic waves passing through coil 10 inthe optical path, because of rotation, will vary relatively slowlycompared with the phase difference changes due to modulator 19. Anyphase differences due to rotation, or the Sagnac effect, will merelyshift the phase differences between the two electromagnetic waves. Theamplitude scaling factor of the modulation frequency component of theoutput signal of photodetection system 14, appearing at the output offilter 22, is expected to be set by the sine of this phase differencemodified further only by the factors of a) the amplitude value of thephase modulation of these waves due to modulator 19 and generator 20,and b) a constant representing the various gains through the system.Then, the periodic effects of this sinusoidal modulation due togenerator 20 and modulator 19 in this signal component are expected tobe removed by demodulation in the system containing phase-sensitivedetector 23 leaving a demodulator system (detector) output signaldepending on just the amplitude scaling factor thereof.

Thus, the voltage at the output of amplifier 21 will typically appearas:

    v.sub.21-out =k{1+cos[φ.sub.R +φ.sub.m cos(ω.sub.g t+θ)]}

The constant k represents the gains through the system to the output ofamplifier 21. The symbol, θ, represents additional phase delay in theoutput signal of amplifier 21 with respect to the phase of the signalprovided by generator 20. Some of this phase shift will be introduced inphotodetection system 14, and some will be due from other sources suchas a phase shift across modulator 19 between the phase of the signalssupplied by generator 20 and the response of modulator 19 in having theindex of refraction of the media therein, or its length, correspondinglychange. The other symbols used in the preceding equation have the samemeaning as they did in the first equation above.

The foregoing equation can be expanded in a Bessel series expansion togive the following: ##EQU2## This signal at the output of amplifier 21is applied to the input of filter 22.

Filter 22, as indicated above, passes primarily the first harmonic fromthe last equation, i.e. the modulation frequency component. As a result,the output signal of filter 22 can be written as follows:

    v.sub.22-out =-2kJ.sub.1 (φ.sub.m)sinφ.sub.R cos(ω.sub.g t+θ+φ.sub.1)

The further phase delay term appearing, Ψ₁, is the additional phaseshift in the first harmonic term added as a result of passing throughfilter 22. This added phase shift is expected to be substantiallyconstant and a known characteristic of filter 22.

The signal from filter 22 is then applied to phase-sensitive detector23, as is the signal from bias modulator generator 20, the latter againbeing equal to C₁ sin(ω_(g) t) where ω_(g) is the radian frequencyequivalent of the modulation frequency f_(g). Assuming that a phaseshift equal to θ+Ψ₁ can be added by phase-sensitive detector 23 to itsoutput signal, the output of that detector will then be the following:

    v.sub.23-out =k'J.sub.1 (φ.sub.m)sinφ.sub.R

The constant k' accounts for the system gains through phase-sensitivedetector 23.

As can be seen from this last equation, the output of phase-sensitivedetector 23 depends on the amplitude φ_(m) supplied by bias modulator 19as operated by bias modulation generator 20. Hence, the amplitude of thesignals supplied by bias modulation generator 20 can be used to set thevalue of the signal at the output of phase-sensitive detector 23 for agiven rotation rate of coil 10 about its axis, i.e. set the scale factorfor the gyroscope at least within a range of possible values therefor.

There are several reasons why an operator of a fiber optic gyroscopewould like to be able to set the amplitude of the bias phase modulationin the system of FIG. 1 to a selected value. That amplitude affects thedistortion which results in the optical waves traveling in the opticalfiber, as well as the noise effectively generated by bias modulationgenerator 20 through its inducing of signals in other parts of theelectronic portion of the system. In addition, of course, the signalstrength at the photodetector output is obviously determined withinlimits by the phase modulation amplitude.

Furthermore, once the phase modulation amplitude is chosen, there isstrong need to maintain that value chosen for the scale factor as aconstant. The scale factor of the fiber optic gyroscope is what will beused by the systems which receive the rotation sensor output signal todetermine what rotation rate is being represented by that signal. Thus,unanticipated changes in that scale factor value will lead to errors inthe value of the angular rotation information being supplied to theseother systems. In those fiber optic gyroscopes which have relativelyundemanding scale factor selection and stability requirements,selecting, and then maintaining, a stable amplitude of the signalsupplied by bias modulator generator 20 would be sufficient. However,fiber optic gyroscopes are very often required to meet much moredemanding requirements with respect to selecting scale factor values,and with respect to stability insofar as maintaining the scale factorselected. Also, there could occur a sufficiently large disturbance inthe phase modulation amplitude such that the scale factor selecting andstabilizing arrangement results in a scale factor having an undesiredvalue. Thus, there is desired a scale factor selecting and stabilizingarrangement which will permit the selection of scale factors from abroad continuous range and the stable maintenance thereof once selected.

SUMMARY OF THE INVENTION

The present invention provides for a rotation sensor, having at least inpart the configuration described above, to have additionally a biasmodulation amplitude monitoring system with first and secondphotodetector output signal component determination arrangements thatcan provide the magnitudes of first and second frequency components inthe photodetector output signal, and a component relationshipdetermination arrangement capable of adjusting the photodetector outputsignal so that the rotation sensor has a selected value in a range ofvalues for a selected rate of its rotation, and so that deviationstherein due to any phase modulation amplitude variations are corrected,based on the relative values of these first and second frequency comagnitudes. This component relationship d arrangement can have a signalprocessor therein which receives the signals representing said first andsecond frequency component magnitudes, and receives the photodetectoroutput signal, to provide a rotation sensor output signal based on thephotodetector output signal after adjustment thereof based in turn onthe relative magnitudes of the first and second frequency componentmagnitudes. Alternatively, the component relationship determinationarrangement can be connected to the optical modulator in a manner so asto be able to control its amplitude, this component relationshipdetermination arrangement in this situation being capable of providing asignal at its output based on the relative said first and secondfrequency component magnitudes to control the optical phase modulator. Areset arrangement is provided capable of resetting the componentrelationship determination arrangement if a sufficiently largedisturbance occurs in the phase modulation amplitude.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a mixed block and circuit schematic diagram of a systemknown in the prior art,

FIG. 2, shows a mixed block and circuit schematic diagram embodying aportion of the present invention which can be substituted for a portionof the diagram of FIG. 1,

FIG. 3 shows a graph representing the behavior of a parametercharacterizing the present invention,

FIG. 4 shows a mixed block and circuit schematic diagram embodying aportion of the present invention which can be substituted for a portionof the diagram of FIG. 1,

FIG. 5 shows a mixed block and circuit schematic diagram embodying thepresent invention which ca be substituted for a portion of the diagramof FIG. 1,

FIG. 6 shows a mixed block and circuit schematic diagram embodying thepresent invention which can be substituted for a portion of the diagramof FIG. 1,

FIG. 7 shows a mixed block and circuit schematic diagram embodying thepresent invention which can be substituted for a portion of the diagramof FIG. 1, and

FIG. 8 shows a mixed block and circuit schematic diagram embodying thepresent invention which can be substituted for a portion of the diagramof FIG. 1.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Because of the difficulty in relying on the output amplitude setting ofbias modulator signal generator 20 to select and maintain an accuratescale factor for the fiber optic gyroscope of FIG. 1, an alternativesystem embodying the present invention is shown in FIG. 2 for thispurpose. FIG. 2 shows a feedback arrangement for controlling theamplitude of the signal used to operate optical phase modulator 19. Thesystem of FIG. 2 permits not only maintaining accurately the scalefactor relating the signal at the output of phase sensitive detector 23to the rotation rate about the axis of coiled optical fiber 10, but alsopermits selecting the value of the scale factor to be so maintained froma substantial range of values. The same designations are used in FIG. 2as were used in FIG. 1 for similar components there.

The input signal for the new portion of the feedback loop is the outputof amplifier 21, given above as the signal v_(21-out) in both closedform and in expanded form. This signal is applied to a furtheramplifier, 24, serving as a buffer to avoid any loading of amplifier 21.In providing this service, amplifier 24 does not alter in anysignificant way the signal provided at the output of amplifier 21.

The signal at the output of amplifier 24 is concurrently supplied to twofilters, 25 and 26. Filter 25 is a high-pass filter which blocks passageof that component of the photodetector output signal from photodetectionsystem 14, after passing through amplifiers 21 and 24, which does notdepend on the frequency ω_(g) of bias modulation signal generator 20.That is, the lowest frequency content signal component of the signal atthe output of amplifier 24, represented by the first term in the aboveexpansion for v_(21-out), is blocked by filter 25.

Rather than a high-pass filter, filter 25 can be a bandpass filterhaving a bandwidth which goes substantially beyond the frequency valueof the signal component chosen from among the remaining signalcomponents in the output signal of amplifier 24 to be used in thefollowing portions of the system of FIG. 2. Further, if the chosenharmonic component is large with respect to other harmonic components,filter 25 may not be necessary in some circumstances.

The system in FIG. 2 will be described on the assumption that the secondharmonic component of the output signal of photodetection system 14, asrepresented at the output of amplifier 24, has been chosen for such usebut other even harmonic components could alternatively be used such thefourth harmonic. In any of these even harmonic situations, the uppercutoff frequency of filter 25, if a bandpass filter, must besubstantially in excess of the harmonic component chosen to besubsequently used to avoid causing any undue phase delay problems withrespect thereto. The phase of such a frequency component must not bedelayed significantly through filter 25 if required frequency conversionof that component to obtain a signal representing its amplitude is to besuccessfully performed.

Filter 26, on the other hand, is a low-pass filter which is used toextract the component of the photodetector output signal provided byphotodetection system 14, as it appears at the output of amplifier 24,which does not have any dependence on the frequency ω_(g) of the biasmodulation signal generator, i.e. the signal component having the lowestfrequency content. This is, as previously indicated, the first term inthe above expansion for the output voltage signal of amplifier 21, afteramplification by amplifier 24, and can be written as follows:

    v.sub.26-out =K.sub.A [1+J.sub.o (φ.sub.m)cosφ.sub.R ]cosθ

The constant k_(A) represents the combined gains of amplifier 21 and 24a well as the intensity of the electromagnetic waves incident onphotodetector 13. The cutoff frequency for low-pass filter 26 is chosento be of relatively low value since this output signal will typicallyhave a low frequency content and be a constant in the absence of anychanging angular rotation rate of the gyroscope. Typical values of thiscutoff frequency would be between 5 and 100 Hz. The phase delay throughthe filter will make no significant difference, and so no special effortis required to control the phase delay added by filter 26. This beingso, that phase delay is not represented in the foregoing equation.

The output of high-pass, or bandpass, filter 25 is provided to afrequency down converter, 27. Frequency down converter 27 also receivesa reference signal of a frequency equal to that of the even harmoniccomponent of the output signal of amplifier 24 chosen to be used in theremaining portions of the system, this reference signal of even harmonicfrequency being based on the frequency of the bias modulation signalgenerator. Since the second harmonic is being used in the presentdescription, a signal with frequency 2ω_(g) is supplied by the biasmodulation signal generator to frequency down converter 27. Thus, thissignal generator is redesignated 20' in FIG. 2 since it now supplies notonly the bias modulation signal at frequency ω_(g), but also the secondharmonic of that signal at frequency 2ω_(g). The resulting signal at theoutput of frequency down converter 27 can be written in the followingmanner:

    v.sub.27-out =k.sub.D k.sub.A J.sub.2 (φ.sub.m)cosθcosφ.sub.R

The constant k_(A) has the same meaning here as it did above. Theconstant k_(D) represents the amplitude value adjustment occurringthrough frequency down converter 27.

These last two signals, v_(26-out) and v_(27-out), representing theamplitudes of selected frequency components of the amplified outputsignal of photodetector 13 obtained from photodetection system 14, areeach scaled by a corresponding selected multiplicative constant beforebeing subtracted one from the other. Thus, the signal resulting from theoutput of high-pass, or bandpass, filter 25 after being converted byfrequency down converter 27, v_(27-out), is multiplied by a selectablemultiplicative constant G₁, and the signal resulting from the low-passfilter 26 is multiplied by the selectable constant G₂.

This latter multiplication is represented by a multiplicative gainblock, 28, in FIG. 2 with the constant G₂ shown therein. Such amultiplication yields the signal

    v.sub.28-out =G.sub.2 k.sub.A [1+J.sub.o (φ.sub.m)cosφ.sub.R ]cosθ

The former multiplication is represented in FIG. 2 by a furthermultiplicative gain block, 29, again with the multiplicative constant G₁shown therein. This results in a signal at the output of that block ofthe form

    v.sub.29-out =G.sub.1 k.sub.D k.sub.A J.sub.2 (φ.sub.m)cosθcosφ.sub.R

This latter signal then has subtracted from it the preceding signal toprovide the following output signal at the output of an algebraicsummer, 30: ##EQU3##

This, difference signal is then integrated over time by an integratorrepresented in FIG. 2 by a block, 31. The result of this integration isapplied to a variable gain amplifier represented in FIG. 2 by a furtherblock, 32. The gain setting of variable gain amplifier 32 determines theamplitude of the bias modulation signal at the bias modulation frequencyω_(g) supplied from bias modulation signal generator 20' to opticalphase modulator 19. The resulting phase modulation of theelectromagnetic waves in the optical system including coiled opticalfiber 10 of FIG. 1 is given effect in the output signal of photodetector13 by these waves impinging thereon to close the feedback loop.

So long as any net signal is provided to integrator 31, that integratorwill integrate the signal over time causing a continually changing valueat its output. Thus, once a value φ_(m).sbsb.o has been selected for theamplitude value φ_(m) of the bias modulation phase shift amplitude andthe system of FIG. 2 settled in a steady state about that value, thesignal value at the output of integrator 31 should be unchanging tothereby cause variable gain amplifier 32 to present at its output asignal, c₁ 'sin(ω_(g) t), which is a suitably multiplied version of theoutput signal c₁ sin(ω_(g) t) of bias modulation signal generator 20',so as to force optical phase modulator 19 to provide just that value ofthe bias phase modulation phase shift amplitude. Thus, this lastdifference signal must be equal to zero when the bias phase modulationphase shift amplitude φ_(m) has a selected value of φ_(m).sbsb.o, or

    v.sub.30-OUT =0=k.sub.A {G.sub.2 [1+J.sub.o (φ.sub.m.sbsb.o)cosφ.sub.R ]-G.sub.1 k.sub.D J.sub.2 (φ.sub.m.sbsb.o)cosφ.sub.R }cosθ

yielding ##EQU4## This last equation, then, is the condition which mustbe met to select a particular value, φ_(m).sbsb.o, for the bias phasemodulation phase shift amplitude φ_(m). Clearly, the value to beselected for the bias phase modulation phase shift amplitude is chosenby the choice of values selected for the constants G₁ and G₂ at leastfor sufficiently slow rotation rates reflected in a sufficiently smallcorresponding Sagnac phase shift φ_(R).

However, disturbances in the value of bias phase modulation amplitudeφ_(m) may occur during operation of the system forcing the feedback loopof FIG. 2 away from steady-state operation because of temperaturechanges, component aging or the like. Such a disturbance in the value ofthe bias phase modulation phase shift amplitude φ_(m) from its desiredvalue φ_(m).sbsb.o can be represented as a small increment or decrementtherefrom, or

    φ.sub.m =φ.sub.m.sbsb.o +δ(t)

where δ(t) represents the small change due to the disturbance from theselected value of the bias phase modulation phase shift amplitudeφ_(m).sbsb.o.

The occurrence of such a disturbance in the value of the bias phasemodulation phase shift amplitude from its selected value of φ_(m).sbsb.oresults in the output signal voltage of summer 30 no longer being zero,and its resulting value can be written as follows:

    v.sub.30-out =k.sub.A {G.sub.2 [1+J.sub.o (φ.sub.m.sbsb.o +δ)cosφ.sub.R ]-G.sub.1 k.sub.D J.sub.2 (φ.sub.m.sbsb.o +δ)cosφ.sub.R }cosθ

If δ is sufficiently small, this last equation may be represented by alinear approximation as follows: ##EQU5## The constant G₂ can beeliminated in the last equation by substituting for it the equationrepresenting the condition to be met for choosing the value of the biasphase modulation phase shift amplitude given above, or ##EQU6##

This signal is then integrated by integrator 31 to provide the followingoutput signal therefrom to control the gain of variable gain amplifier32, this signal being ##EQU7## If a constant K is defined as

    KΔk.sub.I k.sub.A G.sub.1 k.sub.D cosθ

the last equation can be simplified, assuming sufficiently low rotationrates so that cos φ_(R) approximately equals 1, to give ##EQU8## Thelast defined equation has a term therein K' dependent on the phasemodulation phase shift amplitude φ_(m) and the Sagnac phase changeinduced by the rotation of the sensor φ_(R) which, if sufficientlysmall, will leave this factor approximately a constant having a valuedepending on the value φ_(m).sbsb.o selected for the bias phasemodulation phase shift amplitude of the system by the choice of valuesfor the constants G₁ and G₂.

A graph of the factor K' as a function of φ_(m).sbsb.o is shown in FIG.3 (for cosφ_(R) about 1, otherwise there would be a family curve in FIG.3 for different values of φ_(R)). Since the signal v_(31-out) at theoutput of the integrator is the signal which controls the feedback thatdetermines the change in the value of φ_(m) through controlling, throughvariable gain amplifier 32, the amplitude of the signal from biasmodulation generator 20' applied to phase optical modulator 19, thegraph shows that the feedback remains negative in the feedback loop forvalues of φ_(m) which are less than approximately 3.4 to 3.5 radians.Thus, the feedback loop will be stable and act to damp out suchdisturbances for such values of φ_(m) in the bias phase modulation phaseshift amplitude between the waves propagating in opposite directions infiber optic coil 10 so long as the rotation rates of the sensor aboutthe axis of that coil are sufficiently low.

This damping of a disturbance can be shown by the effect of the signalat the output of integrator 31 upon variable gain amplifier 32 andoptical phase modulator 19. Thus, assuming that bias modulationgenerator 20' provides a substantially constant amplitude in its outputsignal v_(32-out), the output signal of variable gain amplifier 32 is

    v.sub.32-out =(c.sub.2 +c.sub.3 v.sub.31-out)v.sub.20'-out

This signal can be rewritten in another form based on there being asignal provided by variable gain amplifier 32 which has a maximumamplitude, V_(VGA).sbsb.o, corresponding to there being no disturbancein the value of the bias phase modulation phase shift amplitude, i.e.that this latter amplitude is at its desired value of φ_(m).sbsb.o, withan increment or a decrement due to the disturbance which, forsimplicity, will again be termed v_(31-out) thereby ignoring its steadystate component, to provide the following alternative representation ofvariable gain amplifier 32

    v.sub.32-out =V.sub.VGA.sbsb.o +k.sub.VGA v.sub.31-out

The constant k_(VGA) represents as a constant the gain effect ofvariable gain amplifier 32. Optical phase modulator 19 will also beassumed to be linear so that the output phase shift it provides can berepresented as

    φ.sub.m =k.sub.OPM v.sub.32-out

The constant k_(OPM) represents as a constant the gain effect of opticalphase modulator 19.

As a result, the value of the optical phase modulator phase shiftamplitude, in a form reflecting a disturbance of magnitude δ, can berelated to the signal at the output of integrator 31. Thus, ##EQU9##using the value for the output signal v_(31-out) of integrator 31 foundabove. Differentiating this last equation with respect to time gives thefollowing result: ##EQU10## The first term on the right of the equalsign will be taken to yield a zero value since any change in k_(OPM)V_(VGA).sbsb.o will be taken to lead to the occurrence of thedisturbance δ(t), and so represented by it. If that is done, and thedisturbance is arbitrarily assumed to have occurred at time t=0, thefollowing first order differential equation results ##EQU11## which iseasily solved to provide a solution of the following form:

    δ(t)=δ.sub.o e.sup.k.sbsp.OPM.sup.k.sbsp.VGB.sup.K't

As indicated above, K' is negative for φ_(m).sbsb.o having a value lessthan approximately 3.4 to 3.5. Thus, the last equation shows that thedisturbance will be damped out.

The output signal of the fiber optic gyroscope provided at the output ofphase sensitive detector 23 was shown above as

    v.sub.32-out =k'J.sub.1 (φ.sub.m)cosθsinφ.sub.R

which, in the presence of a disturbance δ(t), will be

    v.sub.23-out =k'J.sub.1 (φ.sub.m.sbsb.o +δ)cosθsinφ.sub.R

Again, for sufficiently small disturbances, at sufficiently low rotationrates, this can be approximated linearly to result in the followingequation: ##EQU12## As can be seen, there is a resulting error in thescale factor of ##EQU13## Substituting the solution to the first orderdifferential equation above shows that this scale factor error will alsobe damped out with the same time constant controlling the damping of thedisturbance itself: ##EQU14##

Hence, a choice of a value for the bias phase modulation signalamplitude φ_(m).sbsb.o, which is sufficiently smaller than 3.4 to 3.5radians and implemented through corresponding choices for the values ofthe constants G₁ and G₂, will, in the system of FIG. 2, be maintainedagainst disturbances in the value of that choice for sufficiently lowrotation rates of the fiber optic gyroscope about the axis of coiledoptical fiber 10. If φ_(m).sbsb.o must be of a larger value than 3.4 to3.5 radians, an even harmonic greater than the second harmonic must bechosen from the photodetector signal provided by photodetection system14 by filter 25 and frequency down converter 27, such as the fourthharmonic, for use in the system of FIG. 2.

For larger input rates φ_(R), the preceding analysis does not entirelyhold. However, the requirement that V_(30-out) =0 will still be met bythe system of FIG. 2. In this situation, the amplitude φ_(m) will notstay at an initial selected value φ_(m) but will be driven predictablyto another value in unique correspondence with φ_(R) representing thelarger input rate. Thus, the stability of the scale factor of the systemof FIG. 2, relating the system output signal to the input rate φ_(R),will not be significantly reduced.

As an alternative to the feedback arrangement of FIG. 2, the evenharmonic chosen from the photodetector output signal provided byphotodetection system 14, and the lowest frequency component of thatsignal, or the component closest to the zero frequency value, can beused to provide a basis for signal processing calculations toeffectively select a scale factor, and to counter any disturbances inthe fiber optic gyroscope system which would otherwise tend to alter thescale factor so chosen. Such a system is shown in FIG. 4 where the samedesignations are used there that are used in FIGS. 1 and 3 for similarcomponents in each of these figures. Thus, the same equations govern thesignals obtained at the outputs of filter 26 and frequency downconverter 27 as are found for the similar signals in FIG. 2, or

    v.sub.26-out =k.sub.A [1+J.sub.o (φ.sub.m)cosφ.sub.R ]cosθ

    and

    v.sub.27-out =k.sub.A k.sub.D J.sub.2 (φ.sub.m)cosθcosφ.sub.R

Similarly, the same equation characterizes the output of phase sensitivedetector 23 in the system of FIG. 4 as it did in FIG. 2, or

    v.sub.23-out =k'J.sub.1 (φ.sub.m)cosθsinφ.sub.R

As can be seen in FIG. 4, all three of these signals are provided to asignal processor, 40. Such a signal processor can take many forms,perhaps the most straightforward of which would provide a correspondinganalog-to-digital converter for each of these incoming signals, ormultiplex them together through such converter, and transmit theconversion results to the microprocessor. Alternatively, selected ratiosof these signals could be found while in their analog form, and thoseresults submitted to one or more analog-to-digital converters to providethe conversion results to a microprocessor. Another alternative would beto use an existing signal processing integrated circuit chip from thosewhich are currently commercially available.

In any event, one can see from the last three equations that there arethree unknowns therein, φ_(m), φ_(R) and k' assuming that the constantcharacterizing the amplification by amplifiers 21 and 24 and theeffective gain in the conversion in photodetection system 14 fromcurrent to voltage, k_(A), the constant characterizing frequency downconverter 27, k_(D), and phase delay through the optical system θ areknown (or removable in the case of the phase delay). Since there arethree equations for these three unknowns, the values of these threeunknowns can be extracted by signal processor 40 for values of φ_(m)less than about 3.4 to 3.5 radians from which signal processor 40 canprovide an output representing the actual rate of rotation of the fiberoptic gyroscope about the axis of its coiled optical fiber based on thevalue φ_(R). A nominal value for φ_(m), the value of the optical phasemodulation amplitude, can be set by choosing the amplitude of the signalprovided by bias modulation generator 20'.

The fiber optic gyroscopes of FIGS. 2 and 4 are open loop gyroscopes asopposed to closed loop gyroscopes in which the phase differences betweenthe electromagnetic waves propagating in opposite directions are nulledin a feedback loop. However, the even harmonics remain present in theoptical signals so that the system of FIGS. 2 and 4 could be used withclosed loop gyroscopes should there be any reason found for doing so.

As previously described, the desired bias modulation phase shiftamplitude, φ_(m).sbsb.o, is obtained by selecting values for G1 and G2.For each pair of values for G1 and G2, there exists more than oneφ_(m).sbsb.o that will satisfy the equation ##EQU15## However, forpositive values of G2/k_(D) G1, there is only one solution to the aboveequation where φ_(m).sbsb.o is less than π radians, a value having asignificance that will be indicated below. A possible problem in thesystem described above is that, for sufficiently large disturbances inthe values of φ_(m) or φ_(R), the feedback arrangement of FIG. 2 canforce the value of φ_(m) toward a solution φ_(m).sbsb.o to the aboveequation that has a value greater than π. Therefore, an added protectionfor the bias modulation system to assure operation of that system at theintended value φ_(m).sbsb.o for φ_(m) can be provided through theinclusion therein of a reset means. This reset means is to act to resetthe bias phase modulation amplitude feedback control loop operatingpoint if a disturbance in φ_(m) or φ_(R) forces the loop toward a stableoperating point φ_(m).sbsb.o which would have a value greater than π.

One suitable reset means, shown in FIG. 5 comprises a comparator, 42,electrical signal filters, 44 and 46, a resistive voltage divider, 48,and an analog switch, 50, this switch typically constructed using CMOStransistors. The same designations are used in FIG. 5 as were used inFIG. 2 for similar components.

The input signal for the reset means of the present invention is thevoltage signal v_(24-out) at the output of amplifier 24, a substantiallyproportional version of the output signal from photodetection system 14,which is concurrently supplied to filters 44 and 46. Filter 44 is alow-pass filter with a cutoff frequency value of 7 kHz, and is used toobtain a time averaged value of v_(24-out) which does not have anysignificant dependence on the frequency ω_(g) of bias modulation signalgenerator 20'. The output signal of filter 44 is supplied to the inputof voltage divider 48 which provides a reference level voltage of aselected value at its output. This reference level voltage is applied tothe noninverting input of comparator 42.

Filter 46 is also a low-pass filter and is used as a high frequencynoise filter primarily to remove digital switching noise. Filter 46 hasa cutoff frequency value of 200 kHz which leaves present in its outputsignal frequency components at frequency ω_(g), its second and typicallysome higher harmonics. The output signal of filter 46 is applied to theinverting input of comparator 42.

As can be seen, both the inverting and noninverting inputs of comparator42 are provided with input signals that are based on output signalv_(24-out) of amplifier 24. Therefore, the effects of sufficiently slowchanges in v_(24-out), such as any drift in source intensity, will occurproportionately in each such input signal. This prevents theintroduction of a differential signal therebetween due to such slowchanges in v_(24-out) which would adversely affect the comparison of thesignals by comparator 42. Otherwise, the reset means could possiblycause a reset even though φ_(m) does not approach or exceed π, therebycutting off valid data on the rate of rotation in the gyroscope outputsignal.

If the magnitude of the input signal applied to the inverting input ofcomparator 42 during operation becomes less than the selected value ofthe reference level voltage, comparator 42 will rapidly provide anoutput voltage level shift of typically twelve volts at its output witha slew rate of approximately 50 volts per microsecond. This resetvoltage shift is applied to the input of analog switch 50 connectedacross an integration capacitor, 52, between the output and invertinginput of integrator 31. Application of the reset voltage shift willcause switch 50 to close, reaching an "on resistance" value of typically30 ohms within approximately 100 nanoseconds. This will rapidlydischarge integration capacitor 52 to force output signal v_(31-out) ofintegrator 31 to a value near zero, and will do so before the biasmodulation amplitude feedback control loop is driven to an erroneous butstable value for φ_(m).sbsb.o.

When v_(31-out) is zero, integrator 31 and both variable gain amplifier32 and generator 20' will together provide an amplifier output signalhaving an amplitude which leads phase modulator 19 to provide periodicphase changes at frequency ω_(g) with an amplitude that is a fraction ofthe desired value φ_(m).sbsb.o. Thus, optical phase modulator 19 willcontinue to provide bias phase modulation, but with a phase shiftamplitude φ_(m) less than a value giving a stable loop operating point,and so the bias feedback loop will force φ_(m) back to the desired,smallest value for φ_(m).sbsb.o (less than π) after the disturbanceends. During the disturbance, the reset means will continue to reset thefeedback loop operating point by discharging capacitor 52 and keepingthat capacitor shunted by switch 50 for approximately 25 percent of eachreset period set by the second harmonic of the bias modulationfrequency. Although not shown, the reset means will also provide anindication signal during a reset of the feedback loop indicating thatthe data supplied representing the rotation rate is invalid.

The value of the reference level voltage is set by the output signal ofvoltage divider 48, v_(48-out). If the value of the filter 46 outputsignal, v_(46-out), during operation becomes less than the value of thereference level voltage, the reset means will reset the voltage value oncapacitor 52 serving as an error memory in the feedback loop to therebyreset the loop operating point.

The selected value of the reference level voltage is derived from arange of possible trip level voltages for the reset means set by theoperating needs of the optical subsystem. The lowest value of that rangemust be high enough so that the reset means will reset the feedback loopoperating point before φ_(m) attains a value which would lead to astable but unwanted operating point. Oppositely, the highest value ofthe range must be low enough so that the reset means will not reset thefeedback loop operating point when φ_(m) is at or near φ_(m).sbsb.o, andφ_(R) is within its specified range.

The lower trip level value specifying the above range is determined byfinding the corresponding lowest minimum value of the photodiode outputphotocurrent, i_(PD).sbsb.13, and the associated value of φ_(m),ignoring any contribution from φ_(R) to be certain that too great anamplitude for φ_(m) will by itself be sufficient to cause a reset of thefeedback loop operating point. The value of i_(PD).sbsb.13 isrepresented by the equation ##EQU16## as indicated above insofar as theoptical subsystem signal output is concerned. Examination of thisequation shows that i_(PD).sbsb.13 has a lowest minimum value of zeroand that, for a zero value of φ_(R), if φ_(m) reaches a value of π,i_(PD).sbsb.13 reaches a value of zero. Therefore, if the trip level atthe output of voltage divider 48 has a value corresponding toi_(PD).sbsb.13 being greater than zero, φ_(m) cannot reach a valuegreater than π without the reset means causing a reset of the feedbackloop operating point. This prevents the feedback loop from attaining astable but unwanted operating point.

The upper trip level value specifying the above range is determined byfinding the corresponding minimum value of i_(PD).sbsb.13 when φ_(R) isat the highest value within its specified range and φ_(m) is at itsdesired value. Analysis of the extrema of the above equation also showsthat the minimum value of i_(PD).sbsb.13 is represented by the equation##EQU17## In the feedback arrangement of the system described above,φ_(m) is typically desired to be less than 2.0 radians and a typicalmaximum rotation rate to be measured results in φ_(R) being specified tobe at or less than 0.6 radians. Substituting these values into this lastequation provides the result that the higher trip level limit is a valuecorresponding to i_(PD).sbsb.13 having a value of (0.07)I_(o). These twolimit values found for i_(PD).sbsb.13 must be multiplied by the variousgain constants characterizing the electronic systems between the outputcurrent of photodiode 13 and the output voltage of voltage divider 48 togive the lower and upper trip value limits on the range thereof.

Determining the actual lower and upper trip level voltage valuesrequires taking into account additional considerations such as offsetvoltages and noise in the photodiode signal and the signal processingsubsystems acting thereon in providing signals to the reset subsystem.Compensating for these physical limitations requires that the lower andupper trip level voltage limits be altered sufficiently so that thevalues of v_(46-out) which also contains the noise and the offsets,other than those due to divider 48 and the differences between filters44 and 46, are properly positioned with respect to the range of possibletrip values. The reference voltage is selected in the resulting range ofvalues between the upper and lower trip value limits.

A first alternative embodiment of the reset means of the presentinvention is shown in FIG. 6. This reset means is similar to the resetmeans described above except that the signal provided at the input offilter 46 is the output signal of integrator 31 instead of the outputsignal of amplifier 24. The output signal of filter 46 is supplied tothe noninverting input of comparator 42. In addition, resistive voltagedivider 48 will provide a selected reference level voltage, typicallydifferent from that in the reset means described above in being chosento be suitable with the characteristics of the integrator output signal,and supply it to the inverting input of comparator 42. The output signalof integrator 31 is indicative of the magnitude of deviation of φ_(m)from φ_(m).sbsb.o. If the value of this signal is greater than the valueof the predetermined reference level voltage, the reset means will resetthe feedback loop operating point in the same manner as in the resetmeans described above.

In a second alternative embodiment, shown in FIG. 7, the signal at theoutput of amplifier 24 is concurrently supplied to electrical signalfilters 54 and 56. Filter 56 is a high-pass filter with a cutofffrequency value below modulation frequency ω_(g) of bias modulationsignal generator 20'. Filter 56 is used to provide the inverting inputof comparator 51 with the components of photodetector output signalv_(24-out) containing frequencies with values of ω_(g) or its harmonics.Filter 54 is a low-pass filter, also having a cutoff frequency valuebelow ω_(g). Filter 54 is used to provide the noninverting input ofcomparator 51 with the component of the photodetector output signalwhich does not have any significant dependence on ω_(g) or itsharmonics.

For small values of φ_(R), when φ_(m) reaches a value of approximately2.45 radians, the signals at the inverting and noninverting inputs ofcomparator 51 will be equal in value. This results in a reset voltageshift of typically twelve volts being provided at the output ofcomparator 51. The signal at the output of comparator 51 is applied toanalog switch 50 across integration capacitor 52 which resets thefeedback loop operating point in the same manner as in the firstembodiment.

In a third alternative embodiment, shown in FIG. 8, the reset means issimilar to that described in the first embodiment except that the outputsignal of comparator 42 is applied to the input of signal processor 40of FIG. 3. Signal processor 40, upon receiving an indication of a resetvoltage shift, will interrupt its current operation to prevent thedetermination of an erroneous but stable value for φ_(m).sbsb.o. Afterthe disturbance has ended, signal processor 40 will resume itscalculations and again direct generator 20' to provide a generatoroutput amplitude to direct phase modulator 19 to have amplitude φ_(m)equal to the desired value φ_(m).sbsb.o.

Although the present invention has been described with reference topreferred embodiments, workers skilled in the art will recognize thatchanges may be made in form and detail without departing from the spiritand scope of the invention.

What is claimed is:
 1. A bias modulation amplitude monitoring means fora rotation sensor capable of sensing rotation thereof about an axis of acoiled optical fiber to provide a rotation sensor output signalindicative of such rotation through having a pair of electromagneticwaves propagating in said coiled optical fiber in opposite directionsand along other optical path portions to reach and leave said coiledoptical fiber as they travel along an optical path to both impinge on aphotodetector means with a phase difference relationship therebetweenproviding a basis for a resulting photodetector means output signal atan output thereof, and with a bias optical phase modulator positioned ina said optical path portion capable of phase modulating any suchelectromagnetic waves passing therethrough in propagating along saidoptical path so as to provide a varying phase difference between suchelectromagnetic waves of a selectable frequency and a selectableamplitude but with said amplitude subject to variation, said biasmodulation amplitude monitoring means comprising:a first photodetectoroutput signal component determination means having an output and havingan input which is electrically connected to said photodetector meansoutput, said first photodetector output signal component determinationmeans being capable of providing at its output an indication of thatmagnitude occurring in a first component of said photodetector meansoutput signal selected therefrom by its frequency content; a secondphotodetector output signal component determination means having anoutput and having an input which is electrically connected to saidphotodetector means output, said second photodetector output signalcomponent determination means being capable of providing an indicationof that magnitude occurring in a second component of said photodetectormeans output signal selected therefrom by its frequency content; aphotodetector output signal component relationship determination meanshaving a pair of inputs each of which is electrically connected to oneof said first and second photodetector output signal componentdetermination means outputs, said photodetector output signal componentrelationship determination means being capable of adjusting saidphotodetector means output signal such that said rotation sensor outputsignal has a selected value in a range of values for a selected rate ofrotation of said rotation sensor about said axis, and of correctingdeviations therefrom due to said phase modulation amplitude variation,based on relative values of magnitudes of said photodetector means firstand second output signal components; and a reset means having an inputwhich is electrically connected to a selected one of said first andsecond photodetector output signal component determination means, andsaid photodetector output signal component relationship determinationmeans, and having an output which is electrically connected to saidphotodetector output signal component relationship determination means,said reset means being capable of forcing said photodetector outputsignal component relationship determination means to adjust saidphotodetector means output signal such that said varying phasedifference amplitude is reduced to a value below a first predeterminedvalue if, immediately prior to said reduction, a selected second,larger, predetermined value was exceeded by said varying phasedifference amplitude.
 2. The apparatus of claim 1 wherein said resetmeans comprises a decision means, said decision means having an outputwhich is electrically connected to said photodetector output signalcomponent relationship determination means and having both a first and asecond input which are electrically connected to said first and secondphotodetector output signal component determination means, said decisionmeans being capable of providing a reset voltage shift at its output ifthat magnitude of a signal at its second input is less than a referencelevel voltage.
 3. The apparatus of claim 2 wherein said reference levelvoltage is provided to said first input of said decision means by saidphotodetector means output.
 4. The apparatus of claim 2 wherein saidsignal at said second input of said decision means is said photodetectormeans output signal.
 5. The apparatus of claim 2 wherein provision ofsaid reset voltage at said output of said decision means forces saidphotodetector output signal component determination means to adjust saidphotodetector means output signal such that said varying phasedifference amplitude is reduced to a value below said firstpredetermined value.
 6. The apparatus of claim 1 wherein said resetmeans comprises a decision means, said decision means having an outputwhich is electrically connected to said photodetector output signalcomponent relationship determination means and having a first inputwhich is electrically connected to said first and second photodetectoroutput signal component determination means and a second input which iselectrically connected to said photodetector output signal componentrelationship determination means, said decision means being capable ofproviding a reset voltage shift at its output if that magnitude of asignal at its second input is greater than a reference level voltage. 7.The apparatus of claim 6 wherein said reference level voltage isprovided to said first input of said decision means by saidphotodetector means output.
 8. The apparatus of claim 6 whereinprovision of said reset voltage at said output of said decision meansforces said photodetector output signal component determination means toadjust said photodetector means output signal such that said varyingphase difference amplitude is reduced to a value below said firstpredetermined value.
 9. The apparatus of claim 1 wherein said resetmeans comprises a decision means, said decision means having an outputwhich is electrically connected to said photodetector output signalcomponent relationship determination means and having both a first and asecond input that are electrically connected to said first and secondphotodetector output signal component determination means, said decisionmeans being capable of providing a reset voltage shift at its output ifthat magnitude of a signal at its second input is equal to thatmagnitude of a signal at its first input.
 10. The apparatus of claim 9,wherein said signal at said first input of said decision means is onecomponent of said photodetector means output signal and said signal atsaid second input of said decision means comprises those remainingcomponents of said photodetector means output signal.
 11. The apparatusof claim 9 wherein provision of said reset voltage at said output ofsaid decision means forces said photodetector output signal componentdetermination means to adjust said photodetector means output signalsuch that said varying phase difference amplitude is reduced to a valuebelow said first predetermined value.